MAX1951ESA+T - Maxim Integrated

General Description

The MAX1951/MAX1952 high-efficiency, DC-to-DC

step-down switching regulators deliver up to 2A of out-

put current. The devices operate from an input voltage

range of 2.6V to 5.5V and provide an output voltage

from 0.8V to VIN, making the MAX1951/MAX1952 ideal

for on-board postregulation applications. The MAX1951

total output error is less than 1% over load, line, and

temperature.

The MAX1951/MAX1952 operate at a fixed frequency of

1MHz with an efficiency of up to 94%. The high operating

frequency minimizes the size of external components.

Internal soft-start control circuitry reduces inrush current.

Short-circuit and thermal-overload protection improve

design reliability.

The MAX1951 provides an adjustable output from 0.8V

to VIN, whereas the MAX1952 has a preset output of

1.8V. Both devices are available in a space-saving 8-pin

SO package.

Applications

ASIC/DSP/µP/FPGA Core and I/O Voltages

Set-Top Boxes

Cellular Base Stations

Networking and Telecommunications

Features

♦Compact 0.385in2Circuit Footprint

♦10µF Ceramic Input and Output Capacitors, 2µH

Inductor for 1.5A Output

♦Efficiency Up to 94%

♦1% Output Accuracy Over Load, Line, and

Temperature (MAX1951, Up to 1.5A)

♦Guaranteed 2A Output Current

♦Operate from 2.6V to 5.5V Supply

♦Adjustable Output from 0.8V to VIN (MAX1951)

♦Preset Output of 1.8V (1.5% Accuracy) (MAX1952)

♦Internal Digital Soft-Soft

♦Short-Circuit and Thermal-Overload Protection

MAX1951/MAX1952

1MHz, All-Ceramic, 2.6V to 5.5V Input,

2A PWM Step-Down DC-to-DC Regulators

________________________________________________________________ Maxim Integrated Products 1

Ordering Information/

Selector Guide

19-2622; Rev 2; 6/09

For pricing, delivery, and ordering information, please contact Maxim Direct at 1-888-629-4642,

or visit Maxim’s website at www.maxim-ic.com.

PART TEMP RANGE PIN-

PACKAGE OUTPUT

MAX1951ESA+ -40°C to +85°C 8 SO Adj 0.8V to VIN

MAX1952ESA+ -40°C to +85°C 8 SO Fixed 1.8V

MAX1951

OFF

COMP

OUTPUT

0.8V TO VIN, UP TO 2A

INPUT

2.6V TO 5.5V

REF

PGND GND

VCC

OPTIONAL

Typical Operating Circuit

Pin Configuration

PGND

COMPFB

LXREF

GND

VCC

TOP VIEW

MAX1951

MAX1952

+Denotes a lead(Pb)-free/RoHS-compliant package.

MAX1951/MAX1952

1MHz, All-Ceramic, 2.6V to 5.5V Input,

2A PWM Step-Down DC-to-DC Regulators

2 _______________________________________________________________________________________

ABSOLUTE MAXIMUM RATINGS

ELECTRICAL CHARACTERISTICS

(VIN = VCC = 3.3V, PGND = GND, FB in regulation, CREF = 0.1µF, TA= 0°C to +85°C, unless otherwise noted. Typical values are at

TA= +25°C.)

Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional

operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to

absolute maximum rating conditions for extended periods may affect device reliability.

PARAMETER CONDITIONS MIN TYP MAX UNITS

IN AND VCC

IN Voltage Range 2.6 5.5 V

Supply Current S w i tchi ng w i th no l oad , LX unconnected VIN = 5.5V 6 10 mA

Shutdown Current COMP = GND 0.5 1.0 mA

VCC rising 2.35 2.5

VCC Undervoltage Lockout

Threshold When LX starts/stops switching VCC falling 2 2.25 V

REF

REF Voltage IREF = 0µA, VIN = 2.6V to 5.5V 1.96 2 2.03 V

REF Load Regulation IREF = 0 to 40µA, VIN = 2.6V to 5.5V 0.01 0.2 %

REF Line Regulation IREF = 20µA, VIN = 2.6V to 5.5V 0.01 0.4 %

REF Shutdown Resistance From REF to GND, COMP = GND 12 22 Ω

COMP

MAX1951 40 60 80

COMP Transconductance From FB to COMP, VCOMP = 1.25V MAX1952 26.7 40 53.3 µS

COMP Clamp Voltage, Low VIN = 2.6V to 5.5V, VFB = 1.3V 0.6 1 1.2 V

COMP Clamp Voltage, High VIN = 2.6V to 5.5V, VFB = 1.1V 1.97 2.15 2.28 V

COMP Shutdown Resistance From COMP to GND, VIN = 2V 15 30 Ω

COMP rising 0.6 1

COMP Shutdown Threshold When LX starts/stops switching COMP falling 0.17 0.4 V

COMP Startup Current COMP = GND 15 25 40 µA

Output Voltage Range

(MAX1951) When using external feedback resistors to drive FB 0.8 VIN V

VIN = 2.6V to 5.5V MAX1951 0.787 0.795 0.803

FB Regulation Voltage

(Error Amp Only)

C OM P

= 1V to 2V ,

IOU T

= 0 to 1.5A VIN = 2.8V to 5.5V MAX1952 1.773 1.8 1.827 V

FB Input Resistance MAX1952 13 18 28 kΩ

FB Input Bias Current MAX1951 -0.1 +0.1 µA

IN, VCC to GND ........................................................-0.3V to +6V

COMP, FB, REF to GND.............................-0.3V to (VCC + 0.3V)

LX to Current (Note 1).........................................................±4.5A

PGND to GND .............................................Internally Connected

Continuous Power Dissipation (TA = +85°C)

8-Pin SO (derate 12.2mW/°C above +70°C)................976mW

Package Junction-to-Ambient

Thermal Resistance (θJA) (Note 2)...............................82°C/W

Package Junction-to-Case

Thermal Resistance (θJC) (Note 2) ..............................32°C/W

Operating Temperature Range

MAX195_ ESA..................................................-40°C to +85°C

Junction Temperature Range ............................-40°C to +150°C

Storage Temperature Range .............................-65°C to +150°C

Lead Temperature (soldering, 10s) .................................+300°C

Note 1: LX has internal clamp diodes to PGND and IN. Applications that forward bias these diodes should take care not to exceed

the IC’s package power dissipation limits.

Note 2: Package thermal resistances were obtained using the method described in JEDEC specification JESD51-7, using a four-

layer board. For detailed information on package thermal considerations, refer to www.maxim-ic.com/thermal-tutorial.

MAX1951/MAX1952

1MHz, All-Ceramic, 2.6V to 5.5V Input,

2A PWM Step-Down DC-to-DC Regulators

_______________________________________________________________________________________ 3

ELECTRICAL CHARACTERISTICS (continued)

(VIN = VCC = 3.3V, PGND = GND, FB in regulation, CREF = 0.1µF, TA= 0°C to +85°C, unless otherwise noted. Typical values are at

TA= +25°C.)

PARAMETER CONDITIONS MIN TYP MAX UNITS

VIN = 5V 116

VIN = 3.3V 140 266

LX On-Resistance, PMOS

VIN = 2.6V 163

mΩ

VIN = 5V 93

VIN = 3.3V 106 206LX On-Resistance, NMOS

VIN = 2.6V 116

mΩ

LX Current-Sense

Transimpedance From LX to COMP, VIN = 2.6V to 5.5V 0.16 0.24 0.35 Ω

High side 2.4 3.1 4.5

LX Current-Limit Threshold Duty cycle = 100%, VIN = 2.6V to 5.5V Low side -0.6 A

VLX = 5.5V 10

LX Leakage Current VIN = 5.5V LX = GND -10 µA

LX Switching Frequency VIN = 2.6V to 5.5V 0.85 1 1.1 MHz

LX Maximum Duty Cycle VCOMP = 1.5V, LX = Hi-Z, VIN = 2.6V to 5.5V 100 %

LX Minimum Duty Cycle VCOMP = 1V, VIN = 2.6V to 5.5V 15 %

THERMAL CHARACTERISTICS

TJ rising 160

Thermal-Shutdown Threshold When LX starts/stops switching TJ falling 145 °C

ELECTRICAL CHARACTERISTICS

(VIN = VCC = 3.3V, PGND = GND, FB in regulation, CREF = 0.1µF, TA= -40°C to +85°C, unless otherwise noted.) (Note 3)

PARAMETER CONDITIONS MIN TYP MAX UNITS

IN AND VCC

IN Voltage Range 2.6 5.5 V

Supply Current Switching with no load, VIN = 5.5V 10 mA

Shutdown Current COMP = GND 1 mA

VCC rising 2.5

VCC Undervoltage Lockout

Threshold When LX starts/stops switching VCC falling 1.95 V

REF

REF Voltage IREF = 0µA, VIN = 2.6V to 5.5V 1.95 2.03 V

REF Load Regulation IREF = 0 to 40µA, VIN = 2.6V to 5.5V 0.2 %

REF Line Regulation IREF = 20µA, VIN = 2.6V to 5.5V 0.4 %

REF Shutdown Resistance From REF to GND, COMP = GND 22 Ω

MAX1951/MAX1952

1MHz, All-Ceramic, 2.6V to 5.5V Input,

2A PWM Step-Down DC-to-DC Regulators

4 _______________________________________________________________________________________

Note 3: Specifications to -40°C are guaranteed by design and not production tested.

Note 4: The LX output is designed to provide 2.4A RMS current.

ELECTRICAL CHARACTERISTICS (continued)

(VIN = VCC = 3.3V, PGND = GND, FB in regulation, CREF = 0.1µF, TA= -40°C to +85°C, unless otherwise noted.) (Note 3)

PARAMETER CONDITIONS MIN TYP MAX UNITS

COMP

MAX1951 40 80

COMP Transconductance From FB to COMP, VCOMP = 1.25V MAX1952 26.7 53.3 µS

COMP Clamp Voltage, Low VIN = 2.6V to 5.5V, VFB = 1.3V 0.6 1.2 V

COMP Clamp Voltage, High VIN = 2.6V to 5.5V, VFB = 1.1V 1.97 2.28 V

COMP Shutdown Resistance From COMP to GND, VIN = 2V 30 Ω

COMP rising 1.2

COMP Shutdown Threshold When LX starts/stops switching COMP falling 0.17 V

COMP Startup Current COMP = GND 14 40 µA

Output Voltage Range

(MAX1951) When using external feedback resistors to drive FB 0.8 VIN V

MAX1951 0.783 0.807

FB Regulation Voltage

(Error Amp Only) VCOMP = 1V to 2V, VIN = 2.6V to 5.5V MAX1952 1.764 1.836 V

FB Input Resistance From FB to GND MAX1952 10 30 kΩ

FB Input Bias Current MAX1951 -0.1 +0.1 µA

LX (Note 4)

LX On-Resistance, PMOS 266 mΩ

LX On-Resistance, NMOS 206 mΩ

LX Current Sense From LX to COMP, VIN = 2.6V to 5.5V 0.16 0.35 Ω

LX Current-Limit Threshold Duty cycle = 100%, VIN = 2.6V to 5.5V, high side 2.4 4.5 A

VLX = 5.5V 10

LX Leakage Current VIN = 5.5V LX = GND -10 µA

LX Switching Frequency VIN = 2.6V to 5.5V 0.8 1.1 MHz

LX Maximum Duty Cycle VCOMP = 1.5V, LX = Hi-Z, VIN = 2.6V to 5.5V 100 %

MAX1951/MAX1952

1MHz, All-Ceramic, 2.6V to 5.5V Input,

2A PWM Step-Down DC-to-DC Regulators

_______________________________________________________________________________________ 5

EFFICIENCY vs. LOAD CURRENT

(VCC = VIN = 5V)

MAX 1951 toc01

LOAD CURRENT (mA)

EFFICIENCY (%)

1000100

100

10 10,000

VOUT = 3.3V

VOUT = 2.5V

VOUT = 1.5V

VOUT = 0.8V

EFFICIENCY vs. LOAD CURRENT

(VCC = VIN = 3.3V)

MAX 1951 toc02

LOAD CURRENT (mA)

EFFICIENCY (%)

1000100

100

10 10,000

VOUT = 2.5V

VOUT = 1.8V

VOUT = 1.5V

VOUT = 0.8V

REF VOLTAGE

vs. REF OUTPUT CURRENT

MAX1951 toc03

REF OUTPUT CURRENT (μA)

REF VOLTAGE (V)

3530252015105

1.990

1.991

1.992

1.993

1.994

1.995

1.989

040

TA = +85°CTA = +25°C

TA = -40°C

SWITCHING FREQUENCY

vs. INPUT VOLTAGE

MAX1951 toc04

INPUT VOLTAGE (V)

SWITCHING FREQUENCY (MHz)

5.14.63.1 3.6 4.1

0.85

0.90

0.95

1.00

1.05

1.10

1.15

1.20

0.80

2.6 5.6

TA = +85°C

TA = +25°C

TA = -40°C

OUTPUT VOLTAGE DEVIATION

vs. LOAD CURRENT

MAX1951 toc05

LOAD CURRENT (A)

OUTPUT VOLTAGE DEVIATION (mV)

1.20.80.4

-5

-4

-3

-2

-1

-6

01.6

VOUT = 2.5V VOUT = 3.3V

VOUT = 0.8V VOUT = 1.8V

Typical Operating Characteristics

(Typical values are at VIN = VCC = 5V, VOUT = 1.5V, IOUT = 1.5A, and TA= +25°C, unless otherwise noted. See Figure 2.)

LOAD-TRANSIENT RESPONSE

MAX1951 toc06

40μs/div

OUTPUT VOLTAGE:

100mV/div,

AC-COUPLED

OUTPUT CURRENT:

0.5A/div

VIN = 5V

VOUT = 2.5V

IOUT = 0.5 TO 1A

LOAD-TRANSIENT RESPONSE

MAX1951 toc07

40μs/div

OUTPUT VOLTAGE:

100mV/div,

AC-COUPLED

OUTPUT CURRENT:

0.5A/div

VIN = 3.3V

VOUT = 1.5V

IOUT = 0.5 TO 1A

MAX1951/MAX1952

1MHz, All Ceramic, 2.6V to 5.5V Input,

2A PWM Step-Down DC-to-DC Regulators

6 _______________________________________________________________________________________

Typical Operating Characteristics (continued)

(Typical values are at VIN = VCC = 5V, VOUT = 1.5V, IOUT = 1.5A, and TA= +25°C, unless otherwise noted. See Figure 2.)

SHUTDOWN CURRENT

vs. INPUT VOLTAGE

MAX1951 toc12

INPUT VOLTAGE (V)

SHUTDOWN CURRENT (mA)

5.04.54.03.53.0

0.1

0.2

0.3

0.4

0.5

0.6

0.7

0.8

0.9

1.0

2.5 5.5

SWITCHING WAVEFORMS

MAX1951 toc08

200ns/div

INDUCTOR CURRENT

1A/div

VLX

5V/div

OUTPUT VOLTAGE

10mV/div,

AC-COUPLED

VIN = 3.3V

VOUT = 1.8V

ILOAD = 1.5A

SOFT-START WAVEFORMS

MAX1951 toc09

1ms/div

VCOMP

2V/div

OUTPUT VOLTAGE

1V/div

VIN = VCC = 3.3V

VOUT = 2.5V

ILOAD = 1.5A

SOFT-START WAVEFORMS

MAX1951 toc10

1ms/div

VCOMP

2V/div

OUTPUT VOLTAGE

0.5V/div

VIN = VCC = 3.3V

VOUT = 0.8V

SHUTDOWN WAVEFORMS

MAX1951 toc11

20μs/div

0V VCOMP

2V/div

VLX

5V/div

OUTPUT VOLTAGE

1V/div

VIN = VCC = 3.3V

VOUT = 2.5V

ILOAD = 1.5A

MAX1951/MAX1952

1MHz, All-Ceramic, 2.6V to 5.5V Input,

2A PWM Step-Down DC-to-DC Regulators

_______________________________________________________________________________________ 7

Detailed Description

The MAX1951/MAX1952 high-efficiency switching regula-

tors are small, simple, DC-to-DC step-down converters

capable of delivering up to 2A of output current. The

devices operate in pulse-width modulation (PWM) at a

fixed frequency of 1MHz from a 2.6V to 5.5V input voltage

and provide an output voltage from 0.8V to VIN, making

the MAX1951/MAX1952 ideal for on-board postregula-

tion applications. The high switching frequency allows

for the use of smaller external components, and internal

synchronous rectifiers improve efficiency and eliminate

the typical Schottky free-wheeling diode. Using the on-

resistance of the internal high-side MOSFET to sense

switching currents eliminates current-sense resistors,

further improving efficiency and cost. The MAX1951

total output error over load, line, and temperature (0°C

to +85°C) is less than 1%.

Controller Block Function

The MAX1951/MAX1952 step-down converters use a

PWM current-mode control scheme. An open-loop com-

parator compares the integrated voltage-feedback signal

against the sum of the amplified current-sense signal and

the slope compensation ramp. At each rising edge of the

internal clock, the internal high-side MOSFET turns on

until the PWM comparator trips. During this on-time, cur-

rent ramps up through the inductor, sourcing current to

the output and storing energy in the inductor. The current-

mode feedback system regulates the peak inductor cur-

rent as a function of the output voltage error signal. Since

the average inductor current is nearly the same as the

peak inductor current (<30% ripple current), the circuit

acts as a switch-mode transconductance amplifier. To

preserve inner-loop stability and eliminate inductor stair-

casing, a slope-compensation ramp is summed into the

main PWM comparator. During the second half of the

cycle, the internal high-side p-channel MOSFET turns off,

and the internal low-side n-channel MOSFET turns on.

The inductor releases the stored energy as its current

ramps down while still providing current to the output. The

output capacitor stores charge when the inductor current

exceeds the load current, and discharges when the

inductor current is lower, smoothing the voltage across

the load. Under overload conditions, when the inductor

current exceeds the current limit (see the Current Limit

section), the high-side MOSFET does not turn on at the

rising edge of the clock and the low-side MOSFET

remains on to let the inductor current ramp down.

Current Sense

An internal current-sense amplifier produces a current

signal proportional to the voltage generated by the

high-side MOSFET on-resistance and the inductor cur-

rent (RDS(ON) x ILX). The amplified current-sense signal

and the internal slope compensation signal are

summed together into the comparator’s inverting input.

The PWM comparator turns off the internal high-side

MOSFET when this sum exceeds the output from the

voltage-error amplifier.

Current Limit

The internal high-side MOSFET has a current limit of 3.1A

(typ). If the current flowing out of LX exceeds this limit,

the high-side MOSFET turns off and the synchronous

rectifier turns on. This lowers the duty cycle and causes

the output voltage to droop until the current limit is no

longer exceeded. A synchronous rectifier current limit of

-0.6A (typ) protects the device from current flowing into

LX. If the negative current limit is exceeded, the synchro-

nous rectifier turns off, forcing the inductor current to flow

Pin Description

PIN NAME FUNCTION

CC Supply Voltage. Bypass VCC with 0.1µF

capacitor to ground and 10Ω resistor to IN.

2 REF Reference Bypass. Bypass REF with 0.1µF

capacitor to ground.

3 GND Ground

4FB

Feedback Input. Connect FB to the output to

regulate using the internal feedback resistor

string (MAX1952). Connect an external resistor-

divider from the output to FB and GND to set

the output to a voltage between 0.8V and VIN

(MAX1951).

5 COMP

Regulator Compensation. Connect series RC

network from COMP to GND. Pull COMP below

0.17V to shut down the regulator. COMP =

GND when VIN is less than 2.25V (see the

Compensation and Shutdown Mode section)

6 PGND

Power Ground. Internally connected to GND.

Keep power ground and signal ground planes

separate.

7LX

Inductor Connection. Connect an inductor

between LX and the regulator output.

8IN

Power-Supply Voltage. Input voltage range

from 2.6V to 5.5V. Bypass IN with a 10µF (min)

ceramic capacitor to GND and a 10Ω resistor

to VCC.

MAX1951/MAX1952

1MHz, All-Ceramic, 2.6V to 5.5V Input,

2A PWM Step-Down DC-to-DC Regulators

8 _______________________________________________________________________________________

through the high-side MOSFET body diode, back to the

input, until the beginning of the next cycle or until the

inductor current drops to zero. The MAX1951/MAX1952

utilize a pulse-skip mode to prevent overheating during

short-circuit output conditions. The device enters pulse-

skip mode when the FB voltage drops below 300mV, lim-

iting the current to 3A (typ) and reducing power

dissipation. Normal operation resumes upon removal of

the short-circuit condition.

VCC Decoupling

Due to the high switching frequency and tight output

tolerance (1%), decouple VCC with a 0.1µF capacitor

connected from VCC to GND, and a 10Ωresistor con-

nected from VCC to IN. Place the capacitor as close to

VCC as possible.

Soft-Start

The MAX1951/MAX1952 employ digital soft-start circuitry

to reduce supply inrush current during startup conditions.

When the device exits undervoltage lockout (UVLO), shut-

down mode, or restarts following a thermal-overload

event, or the external pulldown on COMP is released, the

digital soft-start circuitry slowly ramps up the voltages at

REF and FB (see the Soft-Start Waveforms in the Typical

Operating Characteristics).

Undervoltage Lockout

If VCC drops below 2.25V, the UVLO circuit inhibits

switching. Once VCC rises above 2.35V, the UVLO

clears, and the soft-start sequence activates.

Compensation

and Shutdown Mode

The output of the internal transconductance voltage

error amplifier connects to COMP. The normal operation

voltage for COMP is 1V to 2.2V. To shut down the

MAX1951/MAX1952, use an NPN bipolar junction

transistor or a very low output capacitance open-drain

MOSFET to pull COMP to GND. Shutdown mode causes

the internal MOSFETs to stop switching, forces LX to a

high-impedance state, and shorts REF to GND.

Release COMP to exit shutdown and initiate the soft-

start sequence.

Thermal-Overload Protection

Thermal-overload protection limits total power dissipation

in the device. When the junction temperature exceeds TJ

= +160°C, a thermal sensor forces the device into shut-

down, allowing the die to cool. The thermal sensor turns

the device on again after the junction temperature cools

by 15°C, resulting in a pulsed output during continuous

overload conditions. Following a thermal-shutdown condi-

tion, the soft-start sequence begins.

MAX1951

VCC

COMP

CURRENT SENSE

SLOPE

COMP

ERROR

SIGNAL

CLOCK

POSITIVE AND NEGATIVE CURRENT LIMITS

GND

REF

PGND

DAC

SOFT-START/

UVLO

REF

PWM

CONTROL

THERMAL

SHUTDOWN

BANDGAP

REF

1.25V

CLAMP

OSC

RAMP GEN

Figure 1. Functional Diagram

MAX1951/MAX1952

1MHz, All-Ceramic, 2.6V to 5.5V Input,

2A PWM Step-Down DC-to-DC Regulators

_______________________________________________________________________________________ 9

Design Procedure

Output Voltage Selection: Adjustable

(MAX1951) or Preset (MAX1952)

The MAX1951 provides an adjustable output voltage

between 0.8V and VIN. Connect FB to output for 0.8V

output. To set the output voltage of the MAX1951 to a

voltage greater than VFB (0.8V typ), connect the output

to FB and GND using a resistive divider, as shown in

Figure 2a. Choose R2 between 2kΩand 20kΩ, and set

R3 according to the following equation:

R3 = R2 x [(VOUT/VFB) – 1]

The MAX1951 PWM circuitry is capable of a stable min-

imum duty cycle of 18%. This limits the minimum output

voltage that can be generated to 0.18 ✕VIN. Instability

may result for VIN/VOUT ratios below 0.18.

The MAX1952 provides a preset output voltage.

Connect the output to FB, as shown in Figure 2b.

Output Inductor Design

Use a 2µH inductor with a minimum 2A-rated DC cur-

rent for most applications. For best efficiency, use an

inductor with a DC resistance of less than 20mΩand a

saturation current greater than 3A (min). See Table 2

for recommended inductors and manufacturers. For

most designs, derive a reasonable inductor value

(LINIT) from the following equation:

LINIT = VOUT x (VIN - VOUT)/(VIN x LIR x IOUT(MAX) x fSW)

where fSW is the switching frequency (1MHz typ) of the

oscillator. Keep the inductor current ripple percentage

LIR between 20% and 40% of the maximum load cur-

rent for the best compromise of cost, size, and perfor-

mance. Calculate the maximum inductor current as:

IL(MAX) = (1 + LIR/2) x IOUT(MAX)

Check the final values of the inductor with the output

ripple voltage requirement. The output ripple voltage is

given by:

VRIPPLE = VOUT x (VIN - VOUT) x ESR / (VIN x LFINAL x fSW)

where ESR is the equivalent series resistance of the

output capacitors.

Input Capacitor Design

The input filter capacitor reduces peak currents drawn

from the power source and reduces noise and voltage

ripple on the input caused by the circuit’s switching.

The input capacitor must meet the ripple current

requirement (IRMS) imposed by the switching currents

defined by the following equation:

For duty ratios less than 0.5, the input capacitor RMS

current is higher than the calculated current. Therefore,

use a +20% margin when calculating the RMS current

at lower duty cycles. Use ceramic capacitors for their

low ESR, equivalent series inductance (ESL), and lower

cost. Choose a capacitor that exhibits less than 10°C

temperature rise at the maximum operating RMS cur-

rent for optimum long-term reliability.

After determining the input capacitor, check the input

ripple voltage due to capacitor discharge when the

high-side MOSFET turns on. Calculate the input ripple

voltage as follows:

VIN_RIPPLE = (IOUT x VOUT)/(fSW x VIN x CIN)

Keep the input ripple voltage less than 3% of the input

voltage.

Output Capacitor Design

The key selection parameters for the output capacitor

are capacitance, ESR, ESL, and the voltage rating

requirements. These affect the overall stability, output

ripple voltage, and transient response of the DC-to-DC

converter. The output ripple occurs due to variations in

the charge stored in the output capacitor, the voltage

drop due to the capacitor’s ESR, and the voltage drop

due to the capacitor’s ESL. Calculate the output voltage

ripple due to the output capacitance, ESR, and ESL as:

VRIPPLE = VRIPPLE(C) + VRIPPLE(ESR) + VRIPPLE(ESL)

where the output ripple due to output capacitance,

ESR, and ESL is:

VRIPPLE(C) = IP-P/(8 x COUT x fSW)

VRIPPLE(ESR) = IP-P x ESR

VRIPPLE(ESL) = (IP-P/tON) x ESL or (IP-P/tOFF) x ESL,

whichever is greater

and IP-P the peak-to-peak inductor current is:

IP-P = [ (VIN - VOUT )/fSW x L) ] x VOUT/VIN

Use these equations for initial capacitor selection, but

determine final values by testing a prototype or evalua-

tion circuit. As a rule, a smaller ripple current results in

less output voltage ripple. Since the inductor ripple

current is a factor of the inductor value, the output

voltage ripple decreases with larger inductance. Use

ceramic capacitors for their low ESR and ESL at the

switching frequency of the converter. The low ESL of

ceramic capacitors makes ripple voltages negligible.

Load transient response depends on the selected

output capacitor. During a load transient, the output

instantly changes by ESR x ILOAD. Before the controller

can respond, the output deviates further, depending on

the inductor and output capacitor values. After a short

time (see the Load Transient Response graph in the

IVIVVV

RMS IN OUT OUT IN OUT

=× −××(/ ) ( ( ))12

MAX1951/MAX1952

1MHz, All-Ceramic, 2.6V to 5.5V Input,

2A PWM Step-Down DC-to-DC Regulators

10 ______________________________________________________________________________________

Typical Operating Characteristics), the controller

responds by regulating the output voltage back to its

nominal state. The controller response time depends on

the closed-loop bandwidth. A higher bandwidth yields

a faster response time, thus preventing the output from

deviating further from its regulating value.

Compensation Design

The double pole formed by the inductor and output

capacitor of most voltage-mode controllers introduces a

large phase shift, that requires an elaborate compensa-

tion network to stabilize the control loop. The MAX1951/

MAX1952 utilize a current-mode control scheme that reg-

ulates the output voltage by forcing the required current

through the external inductor, eliminating the double pole

caused by the inductor and output capacitor, and greatly

simplifying the compensation network. A simple type 1

compensation with single compensation resistor (R1) and

compensation capacitor (C2) creates a stable and high-

bandwidth loop.

An internal transconductance error amplifier compen-

sates the control loop. Connect a series resistor and

capacitor between COMP (the output of the error ampli-

fier) and GND to form a pole-zero pair. The external

inductor, internal current-sensing circuitry, output

capacitor, and the external compensation circuit deter-

mine the loop system stability. Choose the inductor and

output capacitor based on performance, size, and cost.

Additionally, select the compensation resistor and

capacitor to optimize control-loop stability. The compo-

nent values shown in the typical application circuit

(Figure 2) yield stable operation over a broad range of

input-to-output voltages.

The basic regulator loop consists of a power modulator,

an output feedback divider, and an error amplifier. The

power modulator has DC gain set by gmc x RLOAD,

with a pole-zero pair set by RLOAD, the output capaci-

tor (COUT), and its ESR. The following equations define

the power modulator:

Modulator gain:

GMOD = ΔVOUT/ΔVCOMP = gmc x RLOAD

Modulator pole frequency:

fpMOD = 1 / (2 x πx COUT x (RLOAD+ESR))

Modulator zero frequency:

fzESR = 1 /(2 x πx COUT x ESR)

where, RLOAD = VOUT/IOUT(MAX), and gmc = 4.2S.

The feedback divider has a gain of GFB = VFB / VOUT,

where VFB is equal to 0.8V. The transconductance error

amplifier has a DC gain, GEA(DC), of 70dB. The com-

pensation capacitor, C2, and the output resistance of

the error amplifier, ROEA (20MΩ), set the dominant

pole. C2and R1 set a compensation zero. Calculate the

dominant pole frequency as:

fpEA = 1/(2πx CC x ROEA)

Determine the compensation zero frequency is:

fzEA = 1/(2πx CC x RC)

For best stability and response performance, set the

closed-loop unity-gain frequency much higher than the

modulator pole frequency. In addition, set the closed-

loop crossover unity-gain frequency less than, or equal

to, 1/5 of the switching frequency. However, set the

maximum zero crossing frequency to less than 1/3 of

the zero frequency set by the output capacitance and

its ESR when using POSCAP, SPCAP, OSCON, or other

electrolytic capacitors.The loop-gain equation at the

unity-gain frequency is:

GEA(fc) x GMOD(fc) x VFB/VOUT = 1

where GEA(fc)= gmEA x R1, and GMOD(fc) = gmc x

RLOAD x fpMOD/fC, where gmEA = 60µS.

R1calculated as:

R1= VOUT x K/(gmEA x VFB x GMOD(fc))

where K is the correction factor due to the extra phase

introduced by the current loop at high frequencies

(>100kHz). K is related to the value of the output

capacitance (see Table 1 for values of K vs. C). Set the

error-amplifier compensation zero formed by R1and C2

at the modulator pole frequency at maximum load. C2

is calculated as follows:

C2= (VOUT x COUT/(R1 x IOUT(MAX))

As the load current decreases, the modulator pole also

decreases; however, the modulator gain increases

accordingly, resulting in a constant closed-loop unity-

gain frequency. Use the following numerical example to

calculate R1and C2values of the typical application

circuit of Figure 2a.

Table 1. K Value

VOUT = 1.5V

IOUT(MAX) = 1.5A

COUT = 10µF

RESR = 0.010Ω

gmEA = 60µS

DESCRIPTION

COUT (µF) K

10 0.55

22 0.47

V al ues ar e for outp ut i nd uctance fr om 1.2µH

to 2.2µH . D o not use outp ut i nd uctor s l ar g er

than 2.2µH . U se fC

= 200kH z to cal cul ate R1

MAX1951/MAX1952

1MHz, All-Ceramic, 2.6V to 5.5V Input,

2A PWM Step-Down DC-to-DC Regulators

______________________________________________________________________________________ 11

gmc = 4.2S

fSWITCH = 1MHz

RLOAD = VOUT/IOUT(MAX) = 1.5V/1.5 A = 1Ω

fpMOD = [1/(2πx COUT x (RLOAD + RESR)]

= [1/(2 x π×10 ×10-6 x (1 + 0.01)] = 15.76kHz.

fzESR = [1/(2πxCOUT RESR)]

= [1/(2 x π×10 ×10-6 ×0.01)] = 1.59MHz.

For 2µH output inductor, pick the closed-loop unity-gain

crossover frequency (fC) at 200kHz. Determine the

power modulator gain at fC:

GMOD(fc)= gmc ×RLOAD ×fpMOD/fC= 4.2 ×1 ×

15.76kHz/200kHz = 0.33

then:

R1= VO x K/(gmEA x VFB x GMOD(fc)) = (1.5 x 0.55)/

(60 ×10-6 ×0.8 ×0.33) ≈51.1kΩ(1%)

C2= (VOUT ×COUT)/(R ×IOUT(max) )

= (1.5 × 10 × 10-6)/(51.1k ×1.5)

≈196pF, choose 220pF, 10%

Applications Information

PCB Layout Considerations

Careful PCB layout is critical to achieve clean and sta-

ble operation. The switching power stage requires par-

ticular attention. Follow these guidelines for good PCB

layout:

1) Place decoupling capacitors as close to the IC as

possible. Keep power ground plane (connected to

PGND) and signal ground plane (connected to

GND) separate.

2) Connect input and output capacitors to the power

ground plane; connect all other capacitors to the

signal ground plane.

3) Keep the high-current paths as short and wide as

possible. Keep the path of switching current (C1 to IN

and C1 to PGND) short. Avoid vias in the switching

paths.

4) If possible, connect IN, LX, and PGND separately to

a large copper area to help cool the IC to further

improve efficiency and long-term reliability.

5) Ensure all feedback connections are short and

direct. Place the feedback resistors as close to the

IC as possible.

6) Route high-speed switching nodes away from sensi-

tive analog areas (FB, COMP).

Thermal Considerations

The MAX1951 uses a fused-lead 8-pin SO package with

a RTHJC rating of 32°C/W. The MAX1951 EV kit layout is

optimized for 1.5A. The typical application circuit shown

in Figure 2c was tested with the existing MAX1951 EV kit

layout at +85°C ambient temperature, and GND lead

temperature was measured at +113°C for a typical

device. The estimated junction temperature was

+138°C. Thermal performance can be further improved

with one of the following options:

1) Increase the copper areas connected to GND, LX,

and IN.

2) Provide thermal vias next to GND and IN, to the

ground plane and power plane on the back side of

PCB, with openings in the solder mask next to the

vias to provide better thermal conduction.

3) Provide forced-air cooling to further reduce case

temperature.

MAX1951/MAX1952

1MHz, All-Ceramic, 2.6V to 5.5V Input,

2A PWM Step-Down DC-to-DC Regulators

12 ______________________________________________________________________________________

MAX1951ESA

OFF

51.1kΩ

10Ω

10kΩ

0.1μF

220pF

0.1μF

16.9kΩ

14.7kΩ

2μH

10μF

COMP

1.5V AT 1.5A

2.6V TO 5.5V

REF

GND

PGND

GND

OPTIONAL

SHUTDOWN

CONTROL

COMPONENT VALUES

OUTPUT

VOLTAGE (V)

0.8

1.5

2.5

3.3

R1 (kΩ)

33.2

51.1

82.5

110

R2 (kΩ)

OPEN

16.9

R3 (kΩ)

SHORT

14.7

C2 (pF)

220

Figure 2a. MAX1951 Adjustable Output Typical Application Circuit

MAX1952ESA-18

OFF

68kΩ

10Ω

10kΩ

0.1μF

220pF

0.1μF

2μH

10μF

COMP

1.8V AT 1.5A

2.6V TO 5.5V

REF

PGND

GND

VCC

OPTIONAL

SHUTDOWN

CONTROL

Figure 2b. MAX1952 Fixed-Output Typical Application Circuit

MAX1951/MAX1952

1MHz, All-Ceramic, 2.6V to 5.5V Input,

2A PWM Step-Down DC-to-DC Regulators

______________________________________________________________________________________ 13

MAX1951ESA

OFF

100kΩ

10Ω

10kΩ

0.1μF

100pF

0.1μF

1.1μH

22μF

10μF

COMP

1.8V, 2A

3.3V ±5%

REF

GND

PGND

GND

VCC

OPTIONAL

SHUTDOWN

CONTROL

L1: TOKO A915AY-1R1M

C1: TAIYO YUDEN JMK316BJ106ML

C4: TAIYO YUDEN JMK325BJ226MM

10kΩ

12.7kΩ

Figure 2c. MAX1951 Typical Application Circuit with 2A Output

MAX1951/MAX1952

1MHz, All-Ceramic, 2.6V to 5.5V Input,

2A PWM Step-Down DC-to-DC Regulators

14 ______________________________________________________________________________________

Table 2. External Components List

C O M PO N EN T ( F I G U R E 2 ) FUNCTION DESCRIPTION

L1 Output inductor

2µH ±20% inductor

Sumida CDRH4D28-1R8 or

Toko A915AY-2R0M

C1 Input filtering capacitor

10µF ±20%, 6.3V X5R capacitor

Taiyo Yuden JMK316BJ106ML or

TDK C3216X5R0J106MT

C2 Compensation capacitor

220pF ±10%, 50V capacitor

Murata GRM39X7R221K050AD or

Taiyo Yuden UMK107CH221KZ

C3 Reference bypass capacitor

0.1µF ±20%, 16V X7R capacitor

Taiyo Yuden EMK107BJ104MA,

TDK C1608X7R1C104K, or

Murata GRM 39X7R104K016AD

C4 Output filtering capacitor

10µF ±20%, 6.3V X5R capacitor

Taiyo Yuden JMK316BJ106ML or

TDK C3216X5R0J106MT

C5 VCC bypass capacitor

0.1µF ±20%, 16V X7R capacitor

Taiyo Yuden EMK107BJ104MA,

TDK C1608X7R1C104K, or

Murata GRM 39X7R104K016AD

R1 Loop compensation resistor Figure 2a

R2 Feedback resistor Figure 2a

R3 Feedback resistor Figure 2a

R4 Bypass resistor 10Ω ±5% resistor

R5 S hutd ow n tr ansi stor b ase cur r ent b i as ( op ti onal ) 10kΩ ±5% resistor

Q1 Shutdown transistor (optional)

NPN bipolar junction transistor

Fairchild MMBT3904

Zetex FMMT413

Table 3. Component Suppliers

MANUFACTURER PHONE FAX

Murata 650-964-6321 650-964-8165

Sumida 847-545-6700 847-545-6720

Taiyo Yuden 800-348-2496 847-925-0899

TDK 847-803-6100 847-803-6296

Toko 1-800-PIK-TOKO 408-943-9790

Chip Information

PROCESS: BiCMOS

Package Information

For the latest package outline information and land patterns, go

to www.maxim-ic.com/packages. Note that a “+”, “#”, or “-” in

the package code indicates RoHS status only. Package draw-

ings may show a different suffix character, but the drawing per-

tains to the package regardless of RoHS status.

PACKAGE TYPE PACKAGE CODE DOCUMENT NO.

8 SO S8-F6 21-0041

MAX1951/MAX1952

1MHz, All-Ceramic, 2.6V to 5.5V Input,

2A PWM Step-Down DC-to-DC Regulators

Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are

implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.

Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 ____________________ 15

Revision History

REVISION

NUMBER

REVISION

DATE DESCRIPTION PAGES

CHANGED

0 10/02 Initial release —

1 8/03 Updated data sheet title, Features, Typical Operating Circuit, Detailed Description

and added Thermal Considerations section. 1–15

2 6/09 Revised Ordering Information, Electrical Characteristics, Typical Operating

Characteristics, Pin Description, and the Compensation Design section. 1–7, 10, 11, 14